The use of diodes for supplying power to electronic components in various types of devices is well known in the art. Some examples are rectifiers in AC to DC, or DC to DC conversion, ORing of two or more DC supplies for reliability or redundancy, and voltage clamping for inductive switching circuits including switching regulators.
In such applications, the function of the diode is to conduct current in the forward direction (from anode to cathode) with minimum voltage drop, and also to minimize current flow in the reverse direction (“leakage” current). To reduce the diode forward voltage drop, metal-semiconductor junction (Schottky) diodes are often used, but their forward voltage drops are still hundreds of millivolts and reverse currents may be unacceptably large, especially at higher reverse blocking voltages. Reducing the forward direction voltage and reverse (leakage) current is the predominant figure of merit for diodes in these applications.
To reduce the diode forward voltage drop, it is known in the art that the two terminal diode can be replaced by an active switching device such as a transistor. But the transistor is of course a three terminal device and the third control terminal must be driven with a signal suitable for the application. For many switching regulator applications, this control signal is just the logical complement of the control signal already used for the main switch, and its use creates a synchronous rectifier. By using this signal, the increase in circuit complexity is minimized.
For many other applications including some switching regulators, no control signal for the three terminal diode device is directly available, and sense circuits monitoring the voltages and currents at the anode and cathode terminals must be added to generate the control signal for the third terminal. When the sense circuits are included within a block having only two external terminals—anode and cathode (plus perhaps a power supply), an active diode is created which can generally be directly substituted for an ordinary diode but with improved characteristics.
Both bipolar junction transistors (BJTs) and metal oxide transistors (MOSFETs) have been utilized in known synchronous rectifier implementations. MOSFETs are typically preferable due to their fast switching speed, as faster switching time consequently leads to improved power efficiency. The BJT has the disadvantage of slow switching speed, especially during turn-off time due to the fact that the BJT has a storage time which increases with the depth of saturation. As such, the MOSFET can typically turn off faster due to the absence of such storage time.
Another reason MOSFETs are typically preferred is that MOSFETs are driven by voltage, and therefore do not require a continuous DC gate current but only a charge and discharge current. In contrast, the BJT requires a DC base current, which is typically supplied by the input voltage supply, rather than from the rectified current. Since one factor in reducing power consumption is to use the lowest possible supply current, this base current used for driving the BJT disadvantageously reduces the overall efficiency of the synchronous rectifier circuit.
In addition, in the saturation region, the base current of the BJT becomes very large due to low saturated current gain “β.” In order to minimize this base current loss, a BJT usually operates in the active region. However, a relatively large voltage VCE is required between the collector and emitter for operating the BJT in this active region. This large VCE voltage leads to a large increase in the synchronous rectifier's forward voltage drop and a significant decrease in efficiency. Nonetheless, it is possible to reduce the VCE voltage across the collector-emitter junction, for example, by using a large area BJT device, but the drawback of this method is an increase in cost, size and circuit complexity, as well as the limitation of operating only in the saturation region. Hence, this conflict between operating in the saturation region, which yields low VCE but high base current, and in the active region, which allows low base current but high VCE, becomes the key focus from the perspective of efficiency where a compromise for moderate efficiency is much needed.
One way to correct this problem is to operate the BJT in the quasi-saturation region, as is disclosed in U.S. Pat. No. 6,563,725 (the '725 patent). As described in the '725 patent, the voltage VCE across the collector-emitter junction is controlled in proportion to the current flowing through the transistor and load, thus establishing a partially ON VCE voltage and giving rise to only a moderate base current and large device area for the BJT.
FIG. 1(a) illustrates the schematic of the synchronous rectifier disclosed in the '725 patent, and FIG. 1(b) illustrates the various junctions formed by the BJT 102 of the synchronous rectifier disclosed in the patent. Referring to FIG. 1(a), the circuit comprises a BJT 102, transconductance amplifier 110 and offset voltage source 116. As shown, the output of amplifier 110 is coupled to the base terminal 108 of BJT 102. The collector terminal 106 of the BJT 102 serves as an anode. The emitter terminal 104 of the BJT 102 serves as a cathode terminal, and is coupled to the load through an inductor. The emitter and collector terminals of the BJT 102 are also coupled to the inverting input 112 and the non-inverting input 114 of the transconductance amplifier 110, respectively. The transconductance amplifier 110 functions to sense the collector-emitter voltage (VCE) of BJT 102 and provide a base drive current (IB) that is essentially proportional to the voltage difference between the positive and negative inputs of the amplifier 110. It is noted that the VCE at which IB=0 is “offset” from VCE=0 by a small positive offset voltage 116. As explained in the '725 patent, the offset is necessary to realize an optimal IB vs. VCE relationships over a broad range of BJT collector currents.
In operation, the BJT 102 is initially “off” when a forward voltage is first applied, but the relatively high VCE causes a high forward IB to flow, which provides a high base drive turn-on pulse until VCE falls to the operating level, i.e., a steady-state condition. This momentary forward voltage is analogous to the “forward recovery” voltage for a P-N junction diode.
During conduction, the proportionality between IB and VCE (including Voffset) provides the optimal base drive current for a given conduction current. The turn-off of BJT 102 is initiated by a reverse base current IB when the VCE falls below Voffset or reverses polarity.
In such a conventional synchronous rectification circuit 100, one potential problem is the emitter-base reverse-breakdown voltage BVeb. During operation, the voltage of the cathode terminal changes from near ground GND to the (positive) supply voltage. As the emitter-base reverse-breakdown voltage BVeb is continually being reduced in the newer semiconductor technologies, it is likely that the emitter-base reverse-breakdown voltage BVeb will be exceeded as the voltage of the output terminal changes to the supply voltage level. As a result, it is possible that voltage device breakdown could occur even before a sufficient voltage level is supplied for operation of the circuit.
Another potential issue is the transient response of the transconductance amplifier 110. Since there is no diode or clamp circuit inserted between the cathode terminal 122 and anode terminal 124, when the current of the synchronous rectifier (SR) becomes forward direction, there is no current path until the transconductance amplifier 110 turns ON. Emitter-base junction of the BJT 102 makes a PN junction as shown in FIG. 1(b), but there is no current supply path to the base 108. As such, the voltage at the cathode terminal 122, or emitter 104, and the base 108 becomes substantially negative with respect to the collector 106 where the voltage can far exceed the value of an ordinary diode voltage. The value of this voltage could be, for example, 10 volts or more. As a result, this large voltage not only impairs power efficiency of the SR circuit but also has the capability to exceed the breakdown voltage of the base-collector junction.
Another known synchronous rectifier circuit is disclosed in U.S. Pat. No. 5,420,532 (the '532 patent). FIG. 2 depicts a schematic diagram of the synchronous rectifier circuit disclosed by the '532 patent, and FIG. 3 illustrates a timing chart associated with the operation of the circuit. The object of this circuit is to efficiently switch the inductive loads using energy conservation techniques including turning off a driving switch by recirculating residual load current and clamping the output so as to substantially decrease the power dissipation during inductive load turn-off.
In operation, the circuit 200 senses the voltage at the LX terminal 220 and controls the MOS transistor 204 in the following manner. Specifically, transistor 204 is controlled in two conditional states: ON and OFF. When the input SW 218 is at a voltage value greater than the MOS threshold above the voltage at 226, transistor 202 conducts and drives the inductive load 214. When the input SW 218 drops to approximately 0V, transistor 202 turns off. Since current cannot instantaneously change through an inductor, inductor 212 experiences a negative flyback according to the equation V=L*(di/dt). Therefore, the voltage at the LX terminal 220 begins to go negative due to the recirculation current IL 222, turning transistor 204 ON. Transistor 204 therefore begins conducting the recirculation current IL 222. Since transistor 204 only allows the voltage at the LX terminal 220 to fall to a certain voltage below ground, defined by its ON resistance RON times the recirculation current IL, where this voltage is significantly smaller than the forward voltage drop across a conventional diode, the power dissipation and therefore the power loss from inductor 212 current turn-off is significantly smaller.
Meanwhile, recirculation current IL 222 decreases according to the time constant of inductive load 212. The voltage at the LX terminal 220, which is determined by the ON resistance RON times the recirculation current IL, approaches the threshold voltage of circuit 200. As the voltage at the LX terminal 220 increases to greater than this threshold voltage, transistor 204 automatically turns OFF, causing the voltage at the LX terminal 220 to decrease again until a diode formed by the back gate and source becomes active. On the other hand, when the voltage at the LX terminal 220 becomes lower than the circuit threshold voltage, transistor 204 turns ON because the voltage across the diode formed by the back gate and source becomes high enough to turn transistor 204 ON again. Nevertheless, power loss can become very large at the transistor 204 while cycling this operation between exceeding above and decreasing below this threshold voltage repeatedly, and can result in oscillation, as shown in “Transistor204/Gate” signal of FIG. 3. Therefore, to avoid this undesired oscillation, additional control functions must be incorporated into this configuration, which results in additional cost and size to the circuit.
Moreover, transistor 204 is fully ON until its gate voltage reaches the turn-OFF threshold voltage. In order to turn OFF, the gate capacitance of transistor 204 needs to be fully discharged. Until the gate voltage reduces to below the MOS threshold voltage, transistor 204 stays ON. If this turn-OFF time is long, the current IL 222 begins to flow in the reverse direction from the load causing a significant power loss to the circuit. One possible way to avoid this reverse current is by increasing the MOS threshold voltage. However, this would cause the turn ON delay time of the synchronous rectifier or transistor 204 to increase, thereby degrading efficiency.
With regard to the other components illustrated in the circuit of FIG. 2, it is noted that as the node Lx goes low, NPN transistor 210 begins conducting due to its base-emitter junction becoming forward biased. Since PMOS transistor 208 is in a current mirror configuration with MOS transistor 206, a current proportional to the current in PMOS transistor 208 will conduct through PMOS transistor 206. The magnitude of the current through MOS transistor 206 will depend upon the (W/L) size ratios of PMOS transistors 206 and 208. The current conducting through PMOS transistor 206 creates a voltage across the resistor 216. When the voltage across resistor 216 increases to a MOS threshold voltage above Lx potential, the rectifier device or transistor 204 begins conducting, thus clamping node Lx to a voltage below ground while recirculating the current remaining in inductor 212.
Thus, as is clear from the foregoing, known prior art for driving transistors as synchronous rectifiers or “active” diodes has significant drawbacks. Accordingly, there is a need for a device that can eliminate the foregoing problems and limitations associated with the prior art devices.